The present invention relates to a switching power supply circuit included in various electronic apparatuses as a power supply.
The present assignee has proposed various power supply circuits including a resonant converter on the primary side thereof. Japanese Patent Laid-open No. 2003-235259 discloses one example of the proposed power supply circuits.
FIG. 12 is a circuit diagram illustrating one example of a switching power supply circuit that includes a resonant converter and is constructed based on any of the inventions that have been filed by the present assignee.
The switching converter in the power supply circuit shown in FIG. 12 has a configuration in which a separately-excited current resonant converter constructed by half-bridge connection is combined with a partial voltage resonant circuit that performs voltage resonant operation only at the time of turn-off in the switching.
In the power supply circuit in FIG. 12, coupled to a commercial alternating-current power supply AC is a common mode noise filter formed of two filter capacitors CL and one common mode choke coil CMC.
As a rectifying and smoothing circuit for producing a DC input voltage from an AC input voltage VAC from the commercial power supply AC, a full-wave rectifier circuit formed of a bridge rectifier circuit Di and a smoothing capacitor Ci are provided downstream from the common mode noise filter.
The rectified output from the bridge rectifier circuit Di is charged in the smoothing capacitor Ci, and thus obtained across the smoothing capacitor Ci is a rectified and smoothed voltage Ei (DC input voltage) with the same level as that of the AC input voltage VAC.
As a current resonant converter that is fed with the DC input voltage and implements switching, a switching circuit system is provided in which two switching elements Q1 and Q2 formed of MOS-FETs are connected to each other by half-bridge connection as shown in the drawing. Damper diodes DD1 and DD2 formed of body diodes are connected in parallel with the channel between the drain and source of the switching elements Q1 and Q2, respectively, with having the anode-to-cathode direction indicated in the drawing.
In addition, a partial resonant capacitor Cp is connected in parallel with the channel between the drain and source of the switching element Q2. The capacitance of the partial resonant capacitor Cp and a leakage inductance L1 of a primary winding N1 form a parallel resonant circuit (partial voltage resonant circuit). This partial voltage resonant circuit allows partial voltage resonant operation in which voltage resonance arises only when the switching elements Q1 and Q2 are turned off.
The power supply circuit is provided with an oscillation and drive circuit 2 formed of e.g. general-purpose ICs in order to switching-drive the switching elements Q1 and Q2. The oscillation and drive circuit 2 includes an oscillation circuit and a drive circuit, and applies a drive signal (gate voltage) with a requisite frequency to the gates of the switching elements Q1 and Q2. Thus, the switching elements Q1 and Q2 implement switching operation so that they are alternately turned on/off with the requisite switching frequency.
An isolation converter transformer PIT (Power Isolation Transformer) transmits switching outputs from the switching elements Q1 and Q2 to the secondary side.
One end of the primary winding N1 in the isolation converter transformer PIT is coupled via a primary-side series resonant capacitor C1 to the connecting node (switching output node) between the source of the switching element Q1 and the drain of the switching element Q2, which allows acquisition of the switching outputs.
The other end of the primary winding N1 is connected to the primary-side ground as shown in the drawing.
The series resonant capacitor C1 and the primary winding N1 are connected in series. The capacitance of the series resonant capacitor C1 and the leakage inductance L1 of the primary winding N1 (series resonant winding) in the isolation converter transformer PIT form a primary-side series resonant circuit that offers current resonant operation as the operation of the switching converter.
According to the above description, the primary-side switching converter shown in FIG. 12 offers current resonant operation by the primary-side series resonant circuit (L1-C1) and partial voltage resonant operation by the above-described partial voltage resonant circuit (Cp//L1).
That is, the power supply circuit shown in the drawing employs a configuration in which a resonant circuit that offers a resonant converter as the primary-side switching converter is combined with another resonant circuit. Hereinafter, such a switching converter is referred to as a complex resonant converter.
The isolation converter transformer PIT is constructed of an EE-core that is formed by combining E-cores composed of a ferrite material for example, although the illustration thereof in this drawing is omitted. Furthermore, the primary winding N1 and a secondary winding N2 are wound around the center magnetic leg of the EE-core, with the winding part being divided into the primary side and secondary side.
In addition, a gap with a gap length of 1.0 mm or smaller is provided in the center leg of the EE-core in the isolation converter transformer PIT, so that a coupling coefficient of about 0.80 to 0.90 is obtained between the primary winding N1 and the secondary winding N2.
In practice, a coupling coefficient k of about 0.85 is obtained under the following conditions: a gap G is about 0.8 mm, and the numbers of turns of the primary winding N1 and the secondary winding N2 are 20 T (turn) and 50 T (25 T+25 T), respectively.
The secondary winding N2 in the isolation converter transformer PIT is provided with a center tap connected to the secondary-side ground as shown in the drawing, which divides the secondary winding N2 into a secondary winding portion N2A and a secondary winding portion N2B. In addition, rectifier diodes Do1 and Do2 are connected in series to the secondary winding portion N2A and the secondary winding portion N2B, respectively, and a smoothing capacitor Co for smoothing a rectified output from the rectifier diodes Do1 and Do2 is provided. Thus, a full-wave center-tap rectifier circuit is achieved.
Accordingly, as a voltage across the smoothing capacitor Co, a secondary-side DC output voltage Eo is obtained that is a DC voltage with the same level as that of an alternating voltage induced in each secondary winding portion. The secondary-side DC output voltage Eo is supplied to a main load (not shown) as a main DC voltage, and is branched and input to a control circuit 1 as a detected voltage for constant-voltage control.
The control circuit 1 outputs to the oscillation and drive circuit 2 a control signal as a voltage or current of which level is varied depending on the level of the secondary-side DC output voltage Eo.
Based on the control signal input from the control circuit 1, the oscillation and drive circuit 2 varies the frequency of an oscillation signal generated by the oscillation circuit in the oscillation and drive circuit 2, to thereby change the frequency of a switching drive signal applied to the gates of the switching elements Q1 and Q2. Thus, the switching frequency is varied. If the switching frequency of the switching elements Q1 and Q2 is thus varied depending on the level of the secondary-side DC output voltage Eo, the resonant impedance of the primary-side series resonant circuit changes and therefore energy is also changed that is transmitted from the primary winding N1, which forms the primary-side series resonant circuit, to the secondary side. Accordingly, the level of the secondary-side DC output voltage Eo is also varied. Thus, constant-voltage control for the secondary-side DC output voltage Eo is achieved.
Hereinafter, such a constant-voltage control method in which the switching frequency is varied for stabilizing the output voltage is referred to as a switching frequency control method.
FIG. 13 is a waveform diagram for showing the operations of major parts in the power supply circuit of FIG. 12. This diagram shows the operations when the load power Po is 200 W and 0 W, respectively, in the circuit shown in FIG. 12. It should be noted that the load power Po of 200 W is the maximum load power (Pomax) of the circuit in FIG. 12, and 0 W is the minimum load power (Pomin).
Furthermore, in FIG. 13, the AC input voltage VAC is kept constant at 100 V as the input voltage condition. As the secondary-side DC output voltage Eo, a voltage of 100 V or larger is produced.
In order to achieve the above-described conditions of the load power, input voltage and secondary-side DC output voltage, elements having the following characteristics are selected as the major parts in the circuit in FIG. 12:                the isolation converter transformer PIT having a gap length of 0.8 mm and a coupling coefficient k of about 0.85        the primary winding N1 of which number of turns is 20 T        the secondary winding N2 of which number of turns is 50 T (25 T+25 T, across the center tap)        the primary-side series resonant capacitor C1 having a capacitance of 0.068 μF and        the partial resonant capacitor Cp having a capacitance of 1000 pF.        
Referring to FIG. 13, a rectangular waveform voltage V1 is a voltage across the switching element Q2, and indicates the on/off timings of the switching element Q2.
The period during which the voltage V1 is at the 0 level corresponds to the on-period during which the switching element Q2 is in the on-state. In this on-period, a switching current IQ2 having the illustrated waveform flows through the switching circuit system composed of the switching element Q2 and the clamp diode DD2. In contrast, the period during which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei corresponds to the period during which the switching element Q2 is in the off-state. In this off-period, the switching current IQ2 is at the 0 level as shown in FIG. 13.
In addition, although not illustrated, a voltage across the other switching element Q1 and a switching current flowing in the other switching circuit system (Q1, DD1) have a waveform obtained by shifting by 180° the phase of the waveform of the voltage V1 and the switching current IQ2, respectively. That is, as described above, the switching elements Q1 and Q2 implement switching operation with timings at which they are alternately turned on/off.
A primary-side series resonant current Io (not shown) flows through the primary-side series resonant circuit (C1-N1(L1)) with having a waveform resulting from synthesis of the waveforms of the switching currents flowing in these switching circuits (Q1, DD1) and (Q2, DD2).
A comparison between the waveform of the voltage V1 when the load power Po is 200 W and that when it is 0 W makes it apparent that the switching frequency of the primary side is controlled so that the switching frequency be lower when the secondary-side DC output voltage Eo is supplied to a heavier load (Po=200 W) than that when it is supplied to a lighter load (Po=0 W). Specifically, in response to lowering of the level of the secondary-side DC output voltage Eo due to a heavy load, the switching frequency is decreased. In contrast, in response to a rise of the level of the secondary-side DC output voltage Eo due to a light load, the switching frequency is increased. Such switching frequency changes indicate the fact that constant-voltage control operation by upper-side control is carried out as the switching frequency control method.
In this power supply circuit, as shown in FIG. 13, the peak level of the switching current IQ2 when the load power Po is 200 W is 5.6 Ap while that when the load power Po is 0 W is 0.8 Ap.
The above-described operation on the primary side induces an alternating voltage V2 having the illustrated waveform in the secondary winding N2 of the isolation converter transformer PIT. In the half cycles during which the alternating voltage V2 is positive, a current flows through the rectifier diode Do1 on the secondary side. In contrast, in the half cycles during which the alternating voltage V2 is negative (i.e., the half cycles during which an alternating voltage excited in the secondary winding portion N2B is positive), a current flows through the rectifier diode Do2. Thus, in the full-wave center-tap rectifier circuit on the secondary side, a rectified output current I2 flowing between the center tap of the secondary winding N2 and the secondary-side ground has a waveform of which peak levels appear with the same cycle as that with which the positive and negative peak levels of the alternating voltage V2 appear as shown in FIG. 13.
The peak level of the alternating voltage V2 is equal to the level of the secondary-side DC output voltage Eo. In FIG. 13, the peak levels of the rectified output current I2 in the respective half cycles are different: 3 Ap and 2 Ap. The reason for this will be described later.
When a resonant converter configuration is adopted that stabilizes the secondary-side DC output voltage with a switching frequency control method like the power supply circuit in FIG. 12, there is a tendency that the variable control range of the switching frequency for stabilization is a comparatively wide range.
This respect will be with reference to FIG. 14. FIG. 14 shows the constant-voltage control characteristic of a conventional power supply circuit that employs a switching frequency control method for stabilizing its output voltage. The characteristic is indicated as the relationship between a switching frequency fs and the level of the secondary-side DC output voltage Eo.
The following description for this diagram is based on the premise that the power supply circuit of FIG. 12 employs a so-called upper-side control method as a switching frequency control method. The term upper-side control refers to a control method in which the switching frequency is changed within a range of frequencies higher than a resonant frequency fo of the primary-side series resonant circuit, and a resonant impedance change arising from the switching frequency change is utilized to control the level of the secondary-side DC output voltage Eo.
In general, the resonant impedance of a series resonant circuit becomes lowest when the frequency is the resonant frequency fo. Accordingly, the relationship between the secondary-side DC output voltage Eo and the switching frequency fs in the upper-side control is as follows: a switching frequency fs closer to the resonant frequency fo leads to a higher level of the secondary-side DC output voltage Eo while one remoter from the resonant frequency fo leads to a lower level.
Therefore, under the condition that the load power Po is constant, the function of level of the secondary-side DC output voltage Eo depending on the switching frequency fs draws a quadratic curve in which its peak appears when the switching frequency fs is the same as the resonant frequency fo of the primary-side series resonant circuit, and the level decreases as the switching frequency fs is remoter from the resonant frequency fo.
In addition, for the same switching frequency fs, the level of the secondary-side DC output voltage Eo is different depending on the load power Po. Specifically, the voltage level when the load power is the maximum load power Pomax is lower by a certain value than that when the load power is the minimum load power Pomin. That is, under the condition that the switching frequency fs is fixed, a heavier load results in a lower level of the secondary-side DC output voltage Eo.
When it is aimed to, under such a characteristic, stabilize the secondary-side DC output voltage Eo at a voltage level tg by upper-side control, the requisite variable range (requisite control range) of the switching frequency is a range indicated by Δfs.
In an actual power supply circuit shown in FIG. 12, for example, constant-voltage control is implemented so that the secondary-side DC output voltage Eo is stabilized at 135 V by a switching frequency control method, under the following conditions: an input variation range of 85 V to 120 V of the AC input voltage VAC as an AC 100 V-system input; and the maximum and minimum load powers Pomax and Pomin of 200 W and 0 W (no load), respectively, of the secondary-side DC output voltage Eo, which is a main DC voltage.
Under these conditions, the variable range of the switching frequency fs required for constant-voltage control in a conventional typical power supply circuit is from about 80 kHz to about 200 kHz or higher, i.e., Δfs is 120 kHz or higher. This range is considerably wide.
As a power supply circuit, a so-called wide-range compatible one is known that can operate compatibly with an AC input voltage range of 85 V to 288 V for example, so that it can be used both in areas employing the input voltage AC 100 V-system such as Japan and the United States and in areas employing the AC 200 V-system such as Europe.
Consideration will be made below as to provision of a wide-range compatible configuration for a conventional power supply circuit that implements switching frequency control, typified by the power supply circuit in FIG. 12.
The wide-range compatible circuit can accept an AC input voltage range of 85 V to 288 V for example as described above. Therefore, compared with a single-range compatible one that can accept either one of the AC 100 V-system and the AC 200 V-system for example, the level variation range of the secondary-side DC output voltage Eo is larger. In order to implement constant-voltage control for the secondary-side DC output voltage Eo of which level variation range is wide due to such a wide AC input voltage range, a wide switching frequency control range is required. If a conventional power supply circuit has a switching frequency control range of 80 kHz to 200 kHz for an AC 100 V-system single range as described above, in order for the power supply circuit to have a wide-range compatible configuration, the switching frequency control range needs to be widened to a range of about 80 kHz to 500 kHz.
However, in a present IC (the oscillation and drive circuit 2) for driving switching elements, the upper limit of a possible drive frequency is about 200 kHz. Even if a switching drive IC capable of driving elements with the above-described high frequency can be formed and mounted, driving of switching elements with such a high frequency leads to a significantly low power conversion efficiency, and therefore this IC is impractical in an actual power supply circuit.
Accordingly, it has been thought that it is very difficult to achieve a wide-range compatible configuration in a conventional power supply circuit only by stabilizing operation by a switching frequency control method.
In addition, if the power supply circuit includes a full-wave center-tap rectifier circuit as its secondary-side rectifier circuit like one shown in FIG. 12, the switching frequency control range is further widened in particular.
In the full-wave center-tap rectifier circuit, the secondary winding N2 is center-tapped, and thus two secondary winding portions (N2A, N2B) are formed. In these two secondary winding portions N2A and N2B, in the periods of half cycles of one polarity (hereinafter, one half cycles) of an alternating voltage excited in the secondary winding N2, a rectified current flows through the secondary winding portion N2A, the rectifier diode Do1, the smoothing capacitor Co, and the secondary winding portion N2A in that order. In contrast, in the periods of half cycles of the other polarity (hereinafter, the other half cycles) of the alternating voltage, a rectified current flows through the secondary winding portion N2B, the rectifier diode Do2, the smoothing capacitor Co, and the secondary winding portion N2B in that order.
That is, in full-wave center-tap rectification, in the periods of one half cycles, a current flows through only one of the two secondary winding portions but does not flow through the other winding portion.
In such full-wave center-tap rectifying operation, a given electrostatic capacitance exists between the secondary winding portion N2A and the secondary winding portion N2B, which are each wound around a bobbin in the isolation converter transformer PIT.
The existence of the interwinding capacitance is equivalent to the state in which a capacitor Cp20 is connected in parallel with the secondary winding N2 on the secondary side of the isolation converter transformer PIT as shown in FIG. 12.
The connecting of the capacitor Cp20 in parallel with the secondary winding N2 results in formation of a parallel resonant circuit also on the secondary side, by the leakage inductance of the secondary winding N2 and the capacitance of the capacitor Cp20.
The capacitance of the capacitor Cp20 is determined depending on the number of strands in a litz line used for the secondary winding N2 and the window area of the bobbin around which the secondary winding N2 is wound. In the circuit of FIG. 12 involving the above-described conditions, this capacitance is about 100 pF to 500 pF, which is minute.
Since the parallel resonant circuit is thus formed also on the secondary side, an actual circuit of FIG. 12 involves, as a constant-voltage characteristic about the secondary-side DC output voltage Eo like that shown in FIG. 14, a characteristic shown in FIG. 15.
Referring to FIG. 15, in addition to a resonant frequency fo1 of the primary-side series resonant circuit, a resonant frequency fo2p of the secondary-side parallel resonant circuit exists since the parallel resonant circuit is formed also on the secondary side as described above.
When the load power is the minimum load power Pomin in particular, the existence of the different two resonant points results in, as a characteristic curve, a bimodal curve like the illustrated curve including two peaks: the peak corresponding to the primary-side resonant frequency fo1 and the peak corresponding to the secondary-side resonant frequency fo2p. 
In this case, since the capacitance of the capacitor Cp20 is comparatively minute as described above, when there is a tendency toward a heavier load and thus the level of the secondary-side DC output voltage Eo is comparatively low, the secondary-side resonant point has no effect on the characteristic curve (see the characteristic curve when the load power is the maximum load power Pomax). However, when there arises a tendency toward a lighter heavy and thus the load state approaches no-load state, the level of the secondary-side DC output voltage Eo tends to sharply rise. This level rise yields a bimodal characteristic curve like the characteristic curve when the load power Po is 0 as the secondary-side resonant point is elicited.
A comparison between the bimodal characteristic curve and the characteristic curve in FIG. 14 when the load power Po is 0 W allows understanding of the tendency that, in the no-load state, the switching frequency in the bimodal curve of FIG. 15 is higher than that in a unimodal curve for the same output voltage level.
In addition, according to this tendency, the bimodal curve in FIG. 15 leads to a wider requisite control range Δfs of the switching frequency as is apparent from a comparison between two Δfs in FIGS. 14 and 15.
FIG. 16 is a diagram showing the variation characteristic of the switching frequency fs as a function of load in the circuit of FIG. 12, which includes a full-wave center-tap rectifier circuit as its secondary-side rectifier circuit.
According to this characteristic diagram, using a full-wave center-tap rectifier circuit leads to the tendency that the switching frequency sharply rises when the load power Po approaches 0 W due to the elicitation of the secondary-side resonant point as described above.
According to experiments, the switching frequency fs was about 75.8 kHz when the load power was the maximum load power Pomax. In contrast, when the load power was the minimum load power Pomin, the switching frequency fs rose to about 172.4 kHz.
As described above, if a full-wave center-tap rectifier circuit is formed on the secondary-side as a conventional power supply circuit configuration, the requisite control range Δfs is further widened since the existence of two resonant points due to the resonant circuits on the primary-side and secondary-side leads to the widening thereof.
This further widening of the requisite control range Δfs makes it almost impossible to achieve a wide-range compatible configuration.
Furthermore, a wide control range of the switching frequency also causes a problem that the high-speed response characteristic of stabilizing the secondary-side DC output voltage Eo is lowered.
Some electronic apparatuses involve operation in which the load condition varies in such a manner to be instantaneously switched between the maximum-load state and the almost-no-load state for example. Such load variation is referred to as switching load. A power supply circuit incorporated in such apparatuses needs to adequately stabilize its secondary-side DC output voltage even in response to the load variation called the switching load.
However, when the power supply circuit involves a wide control range of the switching frequency as described with reference to FIGS. 14 and 15, it takes a comparatively long time period for the circuit to vary its switching frequency to the frequency for achieving a requisite level of the secondary-side DC output voltage in response to load variation like the switching load. That is, an unfavorable result is obtained as the response characteristic of constant-voltage control.
In particular, the power supply circuit shown in FIG. 12 has such a switching frequency characteristic for constant-voltage control that the switching frequency greatly varies within a load power range of 0 W to about 25 W as shown in FIG. 16. This characteristic is disadvantageous as a constant-voltage control response characteristic against the switching load.
In addition, using a full-wave center-tap rectifier circuit as the secondary-side rectifier circuit like the circuit in FIG. 12 also leads to another problem that bias magnetization arises in the isolation converter transformer PIT in particular.
Specifically, of the secondary winding portions N2A and N2B, one winding portion is long while the other is short depending on which of two is first wound around the bobbin in the isolation converter transformer PIT. Due to this winding length difference, the coupling coefficient between the primary winding N1 and the secondary winding portion N2A is different from that between the primary winding N1 and the secondary winding portion N2B.
In an actual circuit of FIG. 12, the coupling coefficient k between the primary winding N1 and the secondary winding portion N2A is 0.86 while that between the primary winding N1 and the secondary winding portion N2B is 0.85. Thus, a difference arises between the leakage inductances of the respective winding portions. As a result, as shown in the waveform diagram of FIG. 13, the rectified output current I2 has a waveform in which peak levels in the respective half cycles are different from each other.
Since the peak levels of the rectified current in the respective half cycles are thus different, the peak levels of currents flowing through the rectifier diodes (Do1, Do2) on the secondary side are also different. As a result, the breakdown current level of one rectifier diode needs to be increased compared with the case in which rectified currents having the same peak level flow through the respective diodes. Therefore, a need arises to select a more expensive part with a higher breakdown current level than a part used when rectified currents have the same peak level, which imposes cost-up of fabrication of the power supply circuit.
In addition, the different peak levels of rectified currents also cause a problem that a bias arises between conductive losses in the rectifier diodes Do1 and Do2.
Actual experiments on the power supply circuit shown in FIG. 12 revealed that the AC to DC power conversion efficiency ηAC→DC was about 90.5% when the AC input voltage VAC was 100 V and the load power Po was 200 W. Moreover, the experiments offered a result that an AC input power Pin was about 2.6 W when the load power Po was 0 W.